Method and device for determining the rotational speed of a polyphase machine operating with field orientation and without a sensor

ABSTRACT

The invention relates to A method for determining a rotational velocity of a transducerless polyphase machine that is operated in a field-oriented manner, a stator-current model space vector and a conjugate complex reference space vector being calculated using a complete machine model, and the torques-active and dummy variables being measured from these model space vectors and from the measured stator-current actual space vector, from which active and dummy variables, in each case, a system deviation is determined. These measured system deviations are weighted and finally totaled, this total system deviation being used for adjusting the model rotor angular velocity such that the total system deviation becomes zero. Thus a method for determining the rotational velocity of a transducerless polyphase machine that is operated in a field-oriented manner is possible, breakdown protection being assured.

FIELD OF THE INVENTION

The present invention relates to a method for determining a rotationalvelocity of a transducerless polyphase machine that is operated in afield-oriented manner and to a device for carrying out the method.

BACKGROUND INFORMATION

German Patent Application 195 31 771.8 describes a method in which anestablished stator-current model space vector and a stator-currentactual space vector are, in each case, multiplied by a conjugate complexcomplex reference space vector and the imaginary components of theseproducts are compared with each other. A system deviation occurring inthe imaginary component of the products formed is a measure for thedeviation of the model rotor angular velocity from the actual rotorangular velocity. This system deviation is adjusted to zero using anequalizing controller, so that when the system parameters of the realpolyphase machine and of the machine model agree, the model rotorangular velocity is also equal to the actual rotor angular velocity. Thereal components of these products are compared with each other only inresponse to small stator frequencies, the detected system deviation inthe real component of the products formed being used for readjusting thesystem parameters of the model stator resistance. In this way, a correctreadjustment of the model rotor angular velocity is possible in responseto small stator frequencies of the polyphase machine. In response tothese small stator frequencies, assuming the stator voltage can bepreselected, the currents of the real machine and those of the modelmachine are in practical terms only a function of the parameter ofstator resistance and model stator resistance.

One disadvantage of this conventional method is that in response totorques that are high in the range of the breakdown torque, the torqueand thus the active components of the stator current changes onlyslightly in response to a further increase in the rotor angularfrequency, and when the breakdown point is exceeded, the systemdeviation changes its sign. The consequence of this is that thecontrolled system gain becomes zero and, beyond the breakdown point,even becomes negative, so that a positive feedback arises out of anegative feedback. Thus, using this conventional method and the devicefor carrying out this method, the polyphase machine can no longer beprevented from falling out of synchronism.

A further disadvantage of this method is that, assuming a statorfrequency of zero with an extremely slow change in the rotationalvelocity, the model rotor angular velocity can no longer be determinedusing this conventional method.

In addition, the agreement of the identified rotational velocity withthe machine rotational velocity is decisively a function of the qualityof the machine model that is used. The parameters that are used in themodel must therefore be tracked as a function of the operating andworking point in order to correctly reproduce the relationships in themachine. In this context, in particular the rotor and stator resistancemust be tracked on the basis of changed winding temperatures. Sincethere is not supposed to be a temperature measurement of the respectivewinding, the resistances must be identified on-line. In theaforementioned previous German patent application, a method fordetermining the stator resistance was already indicated, which isparticularly necessary in the case of small stator frequencies. Theinfluence of the stator resistance in the middle and upper frequencyrange is slight.

An identification of the rotor resistance is not possible in roughlystationary operation, since an incorrectly reproduced rotor resistancecannot be distinguished from a mistakenly identified rotational velocityin the stator current. The temperature of the rotor winding cannotsimply be measured using measuring techniques, so that frequently thestator and rotor winding temperatures are equated, which, however, isassured only in the first approach. An incorrect rotor resistancenevertheless leads to an incorrectly identified rotational velocity, sothat this can not be tolerated particularly in operation without arotary transducer.

SUMMARY

The present invention provides a method and device for determining arotational velocity of a transducerless polyphase machine that isoperated in a field-oriented manner such that the aforementioneddisadvantages no longer occur.

As a result of fact that the detected system deviation in the realportion of the products is referred to for identifying the rotationalvelocity, in the range of the breakdown torque, which is usuallyrequired only in the range of high rotational velocities, the controlledsystem gain has its largest value in response to a clearly determinedsign. In the range of small rotor frequencies, there is, however, aclearly reduced controlled system gain. For this reason, the detectedsystem deviations in the imaginary portion and the real portion of theproducts formed are, in each case, weighted and totals using a factor,so that from small to very high rotational velocities, the rotationalvelocity can be correctly identified. Thus a machine's falling out ofsynchronism can at all times be prevented.

In an advantageous method according to the present invention, asupplementary value is added to a total system deviation. Thissupplementary value is determined using a flux magnitude modulation,this value being a function, on the one hand, of the selected amplitudeand the frequency of the flux magnitude modulation and, on the otherhand, decisively of the rotational velocity deviation between the realpolyphase machine and the model machine. The larger the rotationalvelocity error, the larger the value that is added for the total systemdeviation.

Using this advantageous method, it is possible to precisely identifythis rotational velocity irrespective of the quality of the measuringcomponents used, assuming a stator frequency of zero and in response toan extremely slow change of the rotational velocity of the polyphasemachine.

In a further advantageous method according to the present invention, asupplementary value is also formed on the basis of a flux magnitudemodulation, the value being used for adjusting the model rotorresistance. Since the value additionally determined by the fluxmagnitude modulation is not needed for identifying the rotationalvelocity in the middle and upper frequency range, the supplementaryvalue is used in this frequency range for adjusting the model rotorresistance. In this way, the erroneous identification of the rotationalvelocity caused by an incorrectly reproduced rotor resistance iseliminated, so that in this frequency range the rotational velocity isalways correctly identified.

As a result of the method according to the present invention, fordetermining a rotational velocity of a transducerless polyphase machinethat is operated in a field-oriented manner, the rotational velocity ofthe polyphase machine can now always be precisely identified, from astator frequency of zero to very high stator frequencies.

In order to further explain the present invention, reference is made tothe drawing, in which is schematically represented an exemplaryembodiment of a device for carrying out the method according to thepresent invention for determining a rotational velocity of atransducerless polyphase machine that is operated in a field-orientedmanner.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 depicts a basic representation of an induction machine, which issupplied via a pulse-controlled AC converter having essentially constantDC voltage between the input terminals.

FIG. 2 depicts a block diagram of a particularly advantageous device forcarrying out the method according to the present invention.

FIGS. 3 and 4 represent the interrelationships in response to themodulation of the stator flux magnitude.

DETAILED DESCRIPTION

In the following descriptive part, a complex variable is indicated by anunderline, a space vector by an arrow, a model variable by a circumflexaccent, and a conjugate complex variable by an asterisk.

In FIG. 1, a pulse-controlled AC converter SR is symbolicallyrepresented by changeover switches U_(a), U_(b), U_(c). Changeoverswitches U_(a), U_(b), U_(c), as a function of bivalent control signalsS_(a), S_(b), S_(c), connect terminals a, b, and c to positive terminal“+” on the input side of pulse-controlled AC converter SR, if theassociated control signal has the value of 1, and to negative terminal“−”, when the value of the control signal is equal to 0. Measuredagainst median potential M between input terminals “+” and “−”, aterminal acting as a connection to induction machine DM can only acceptvoltage values +E_(d) or −E_(d). As a result of pulse modulation, it ispossible for median value voltages −{overscore (e)}_(aM), −{overscore(e)}_(bM), −{overscore (e)}_(cM), which are formed via a modulationcycle having duration T_(m), to accept each value between +E_(d) and−E_(d). Induction machine DM is symbolically represented by its spacevector-equivalent circuit diagram (Γ space vector equivalent circuitdiagram) in the fixed-stator reference numeral system.

On the basis of the customary simplifications, the following parametersare sufficient for a complete description of the system of the inductionmachine:

Stator resistance R_(S), rotor resistance R_(r), magnetizing inductionL_(μ), leakage inductance L_(σ), as well as electrical angular velocityω of the rotor, it being designated that ω=Ω×p and using Ω as themechanical angular velocity of the rotor relative to the stator andusing p as the number of pole pairs of the machine. Resistanceparameters R_(S) and R_(r) change in accordance with the associatedwinding temperatures and inductances L and Lσ with the flux linkages.

From the variables in phases a, b, and c, represented in the example ofstator currents i_(a), i_(b) and i_(c), the orthogonal coordinates ofthe complex stator-current space vector $\underset{\rightarrow}{I_{s}}$

are obtained in accordance with the following specifications:

i_(Sα)=−i_(b)−i_(c),

$\begin{matrix}{{i_{S\quad \beta} = {\frac{1}{\sqrt{3}}\left( {i_{b} - i_{c}} \right)}}{{\underset{\rightarrow}{i}}_{S} = {i_{\alpha} + {j\quad \cdot i_{\beta}}}}} & (1)\end{matrix}$

where j={square root over (−1+L )}

In order to calculate torque M brought about per pole pair, there aremany possibilities, e.g., $\begin{matrix}{{M = {{{Im} \cdot \underset{\_}{M}} = {\frac{3}{2}\left( {{\underset{\rightarrow}{\Psi}}_{\mu}^{*} \cdot {\underset{\rightarrow}{i}}_{S}} \right)}}}{{where} = {{\underset{\rightarrow}{\Psi}}_{\mu}^{*} = {\Psi_{\mu \quad \alpha} - {j\quad \Psi_{\mu \quad \beta}}}}}} & (2)\end{matrix}$

The real portion of complex variable M, which will be described below,does not contribute anything to the actual torque and should thereforebe designated as dummy (reactive) torque BG.

In place of conjugate complex space vector${\underset{\rightarrow}{\Psi}}_{\mu}^{*}$

of the stator flux linkage, in equation (2) conjugate complex spacevector ${\underset{\rightarrow}{\Psi}}_{r}^{*}$

derived from the rotor flux linkages can also be used, or even spacevector$\left. {{\underset{\rightarrow}{\Psi}}_{r}^{*}/} \middle| {\underset{\rightarrow}{\Psi}}_{r}^{*} \right|,$

the conjugate complex value of the unit space vector in the direction ofrotor flux space vector ${\underset{\rightarrow}{\Psi}}_{r}^{*}.$

All of these products of stator-current space vector${\underset{\rightarrow}{i}}_{s}$

and a complex “transformation variable”${\underset{\rightarrow}{T}}^{*}$

have the property that their imaginary part is proportional to theactual torque, i.e., in response to a stationary symmetrical operationhaving state variables that oscillate in a temporally harmonic manner, atemporally constant zero-frequency quantity is formed. The same appliesto the “dummy variable,” the real part of the products. If, in general,the space vector of the conjugate complex transformation variable isdesignated as${\underset{\rightarrow}{\hat{T}}}^{*} = {{reference}\quad {variable}\quad {for}\quad {splitting}\quad {the}\quad {stator}\quad {current}\quad {into}}$

${\underset{\rightarrow}{i}}_{S\bot} = \text{(torque-)~~active~~component~~of~~the~~~stator~~current~~and}$

${\underset{\rightarrow}{i}}_{S{}} = \text{(torque-)~~dummy~~component~~of~~the~~stator~~current,}$

then $\begin{matrix}{{{\underset{\rightarrow}{\hat{T}}}^{*} \cdot {\underset{\rightarrow}{i}}_{S}} = {{{\underset{\rightarrow}{\hat{T}}}^{*} \cdot {\underset{\rightarrow}{i}}_{S\bot}} + {{\underset{\rightarrow}{\hat{T}}}^{*} \cdot {\underset{\rightarrow}{i}}_{S{}}}}} & (3) \\{{{Im}\left( {{\underset{\rightarrow}{\hat{T}}}^{*} \cdot {\underset{\rightarrow}{i}}_{S}} \right)} = {{{Im}\left( {{\underset{\rightarrow}{\hat{T}}}^{*} \cdot {\underset{\rightarrow}{i}}_{S\bot}} \right)} = {WG}}} & \quad \\\left( {{torques},{{actual}\quad {variable}}} \right) & \quad \\{{{Re}\left( {{\underset{\rightarrow}{\hat{T}}}^{*} \cdot {\underset{\rightarrow}{i}}_{S\bot}} \right)} = {{{Re}\left( {{\underset{\rightarrow}{\hat{T}}}^{*} \cdot {\underset{\rightarrow}{i}}_{S{}}} \right)} = {WG}}} & \quad \\\left( {{torques}\text{-}{dummy}\quad {variable}} \right) & \quad\end{matrix}$

In order to establish whether torques M and M from real machine DM andthe machine model agree, and thus, given otherwise equal systemparameters, also rotational velocities ω and ω agree, it is onlynecessary to multiply the two space vectors of stator currents${\underset{\rightarrow}{i}}_{S}\quad {and}\quad {\hat{\underset{\rightarrow}{i}}}_{S}$

by the same conjugate complex reference variable${\hat{\underset{\rightarrow}{T}}}^{*}$

and to compare the imaginary parts: $\begin{matrix}{{{{Im}\left\lfloor {{\hat{\underset{\rightarrow}{T}}}^{*} \cdot {\hat{\underset{\rightarrow}{i}}}_{S}} \right\rfloor} - {{Im}\left\lfloor {{\hat{\underset{\rightarrow}{T}}}^{*} \cdot {\underset{\rightarrow}{i}}_{S}} \right\rfloor}} = {\Delta\bot}} & (4)\end{matrix}$

If variable Δ⊥ is brought to zero by adjusting model parameter ω, then,in any case, stator-current components${\hat{\underset{\rightarrow}{i}}}_{S\bot}\quad {and}\quad {\underset{\rightarrow}{i}}_{S\bot}$

agree that are orthogonal to complex reference variable${\hat{\underset{\rightarrow}{T}}}^{*};$

for angular velocities {circumflex over (ω)} and ω this applies withcertainty when all other system parameters in real machine DM and in themachine model are also equal.

FIG. 2, for example, shows how a conventional method for controlling andregulating an induction machine DM can be supplemented in order to beable to do without direct measurement of angular velocity Ω in machineDM. In this context, it is assumed that a complete machine modelcontains space vector ${\hat{\underset{\rightarrow}{i}}}_{S}$

of the model stator currents and, in addition, a reference space vectorT̂,

whose direction always corresponds with that of space vector${\hat{\underset{\rightarrow}{\Psi}}}_{\mu},$

of the stator flux linkage or with that of${\hat{\underset{\rightarrow}{\Psi}}}_{r}$

of the rotor flux linkage or of a direction located in between.

A conventional signal processor 2, which includes inter alia amodulator, a closed-loop control, and a complete machine model, normallyrealizes the following functions in regularly sequential calculationcycles having constant duration T_(c):

From setpoint value ŜM for the strength of the magnetic rotating fieldand setpoint value ŜM for the torque, space vector${\overset{\_}{\underset{\rightarrow}{e}}}_{S}$

of phase voltages {overscore (e)}_(aM), {overscore (e)}_(bM), and{overscore (e)}_(cM) are calculated, which are necessary to reduce thedifferences between the setpoint and actual values or to maintain thedifferences at zero, an estimated value {circumflex over (ω)} of therotor angular velocity being necessary.

Based on active input DC voltage 2E_(d) of the power converter SR,measured by measuring element 4, the associated pulse pattern iscalculated for switching variables S_(a), S_(b), and S_(c) the interiorcurrent-dependent voltage drop offs of power converter SR being takeninto account. For this purpose, stator-current space vector${\hat{\underset{\rightarrow}{i}}}_{S}$

of the machine model must first be calculated and then be converted tophase values î_(a), î_(b), and î_(c).

In order to be able to calculate stator-current model space vector${\hat{\underset{\rightarrow}{i}}}_{S}$

and all other necessary or desired variables, the state variables, e.g.,space vectors of stator flux linkage${\hat{\underset{\rightarrow}{\Psi}}}_{\mu}$

and space vector of rotor flux linkage${\hat{\underset{\rightarrow}{\Psi}}}_{r}$

must again be determined in advance in accordance with the controllersof the system. At the end of the calculation cycle having durationT_(c), the pulse pattern of switching variables S_(a), S_(b), S_(c) aredetermined. The associated switching actions are carried out within thenext modulation cycle of total duration T_(m)=N×T_(c), N being a wholenumber, at the previously calculated time points of the semiconductorswitches of power converter SR. At the end of the observed calculationcycle, not only the future value of stator-current model space vector${\hat{\underset{\rightarrow}{i}}}_{S}$

is known at the end of the just begun modulation cycle, but the nextintermediate value of stator-current model space vector${\hat{\underset{\rightarrow}{i}}}_{S},$

which will occur at the end of the next calculation cycle can also bedetermined from the calculated pulse pattern and stored in memory.Associated currents i_(a), i_(b) and i_(c) in real machine DM occur witha delay of T_(c). Until they have been measured by measuring element 6and are available in converted form at resolver 8, usually a furtherdead time having duration T_(c), passes, or a small fraction of thisduration. These effects are symbolically taken into account in FIG. 2through dead time elements 10 and 12. Irrespective of number N ofcalculation cycles, which occur in one modulation cycle, in everycalculation cycle the new value of stator-current actual space vector${\underset{\rightarrow}{i}}_{S},$

of the measured machine currents is compared with associatedstator-current model space vector${\hat{\underset{\rightarrow}{i}}}_{S}$

of the currents of the machine model, which was already calculated andstored in memory a plurality of calculation cycles previously, as issymbolized in FIG. 2 by dead time element 14 having dead time T_(Σ),which is derived from the sum of the other dead times.

If, as reference variable for splitting stator-current space vector${\underset{\rightarrow}{i}}_{S}$

and stator-current model space vector${\hat{\underset{\rightarrow}{i}}}_{S},$

a conjugate complex value of space vector$\hat{\underset{\rightarrow}{T}}$

is used which is chosen as reference variable, the conjugate complexvalue also being situated in the past b (time period) T_(Σ) (dead timeelement 16), in order to form the variables${{\hat{W}G} = {{Im}\left( {{\underset{\rightarrow}{\hat{T}}}^{*} \cdot {\underset{\rightarrow}{\hat{i}}}_{S}} \right)}};{{\hat{B}G} = {{Re}\left( {{\underset{\rightarrow}{\hat{T}}}^{*} \cdot {\hat{\underset{\rightarrow}{i}}}_{S}} \right)}}$and${{{WG} = {{Im}\left( {{\underset{\rightarrow}{\hat{T}}}^{*} \cdot {\underset{\rightarrow}{i}}_{S}} \right)}};{{BG} = {{Re}\left( {{\underset{\rightarrow}{\hat{T}}}^{*} \cdot {\underset{\rightarrow}{i}}_{S}} \right)}}},$

in multiplying elements 18 and 20 in accordance with the above equation(3), then active variables WG and ŴG agree directly with the torques ofreal machine DM and the machine model, down to an identical factor. Foradapting the parameter of rotor velocity ω of the machine model,provision is made for an equalizing controller 22, whose input variable,system deviation Δ⊥ of imaginary parts ŴG and WG of the products formed,is formed in comparator 24. Usually, variables WG and ŴG correspondingto the torques are themselves not temporally constant in stationaryoperation, because the input voltages for the real machine and for themachine model proceed in a pulse-modulated manner. However, if bothvoltages agree in the same way as all system parameters, then, in theequalized state, this system deviation Δ_(⊥) is theoretically equal tozero, practically a close approximation. This means that, for example,amplification and reset time of utilizing controller 22 should beselected such that a clearly better dynamic performance results than isthe case with other methods. In this connection, an essential role isplayed by the measure, according to the present invention, of permittinga dead time to occur in the path of signals${\hat{\underset{\rightarrow}{i}}}_{S}\quad {and}\quad {{\hat{\underset{\rightarrow}{T}}}^{*}.}$

As was already mentioned, the pulse pattern for switching signals S_(a),S_(b), S_(c) is calculated such that, using active input DC voltage2E_(d) and taking into consideration the interior voltage drop offs ofpower converter SR, desired median value${\overset{\_}{\underset{\rightarrow}{e}}}_{S}$

of the machine voltage space vector is obtained for the next modulationcycle. If the precision of this controlling is not sufficient to assurefor real machine DM and the machine model the presupposed equal patternof the space vector of the input voltages, the median values formed viaa modulation cycle of three output voltages {overscore (e)}_(aM),{overscore (e)}_(bM) and {overscore (e)}_(cM) of power converter SR canbe measured via a measuring element 26 and can be used as actual valuesfor three voltage correction closed-loop controls, a dead time, however,occurring in its turn which is taken into account in FIG. 2 by dead timeelement 28.

From another point of view, in order to realize the most rugged methodpossible, it is advantageous that reference variable${\hat{\underset{\rightarrow}{T}}}^{*}$

select the conjugate complex space vector of stator flux linkages${\hat{\underset{\rightarrow}{\Psi}}}_{\mu}^{*}.$

Parameter L_(μ) shows very strong dependence on the flux linkages, butusually only the dependence of fundamental wave inductance L_(μf) on themagnitude of space vector ${\underset{\rightarrow}{\Psi}}_{\mu}$

is taken into account because a precise reproduction of momentarymagnetizing currents i_(μa,b,c) is too costly as a function of themomentary values of the flux linkages. Since magnetizing currentsi_(μa,b,c) do not supply any contribution to the torque, as long as thespace vector of magnetizing currents ${\underset{\rightarrow}{i}}_{\mu}$

remains in the same direction as the space vector${\hat{\underset{\rightarrow}{\Psi}}}_{r}$

of the rotor flux linkage, differences between magnetizing current spacevectors${\underset{\rightarrow}{i}}_{\mu}\quad {and}\quad {\underset{\rightarrow}{i}}_{\mu}$

have no influence on system deviation Δ⊥ of torques-active variables ŴGand WG and thus also not on the time characteristic of the outputvariable of rotor angular velocity ω of equalizing controller 22.

As a result of the measures described above, it is ensured thatstator-current components${\underset{\rightarrow}{i}}_{S\bot}\quad {and}\quad {\underset{\rightarrow}{\hat{i}}}_{S\bot}$

of real polyphase machine DM and the machine model agree, in the dynamicmode considerably and in the stationary mode all but ideally, if that ispossible by adaptation of the parameter of rotor angular velocity ω inthe machine model. In the case of small stator frequencies, with apreselected voltage, the currents of real machine DM and also those ofthe machine model are practically only determined, in the stationarymode, by the parameter of stator resistance R_(S) and {circumflex over(R)}_(S). Stator resistance R_(S) changes very considerably according tothe winding temperature, so that if the model value does not followstator resistances R_(S), system deviation Δ⊥ at the input of equalizingcontroller 22 is not equal to zero, even if model rotor angular velocity{circumflex over (ω)} and rotor angular velocity ω agree. That meansthat a correct determination of model rotor angular velocity {circumflexover (ω)} is then not possible.

Since adjusting the parameter of model rotor angular velocity{circumflex over (ω)} does nothing to change the fact that the voltagesof real machine DM and the machine model agree as far as technicallypossible, system deviation Δ⊥, in response to small stator frequencies,remains different from zero for as long as the differences betweenstator resistance R_(S) and model stator resistance {circumflex over(R)}_(S) remain.

This shortcoming can be removed by comparing dummy torques {circumflexover (B)}G and BG, not previously used, also called torques-dummyvariables. If the parameters of stator resistance R_(S) and model statorresistance {circumflex over (R)}_(S) do not agree, e.g., in response tolow frequencies, then, in response to equal voltages, this leads notonly to differences in torque-forming current components${\underset{\rightarrow}{i}}_{S\bot}\quad {and}\quad {\underset{\rightarrow}{\hat{i}}}_{S\bot}$

of real machine DM and of the machine model, but also to differencesbetween flux-forming current components${\underset{\rightarrow}{i}}_{S{}}\quad {\hat{\underset{\rightarrow}{i}}}_{S{}}$

of real machine DM and the machine model and thus also to inequality ofdummy torques BG and {circumflex over (B)}G. If difference Δ∥ betweenthese dummy torques BG and {circumflex over (B)}G is established in afurther comparator 30, then the parameter of model stator resistance{circumflex over (R)}_(S) can be adjusted via a further equalizingcontroller 32 until this detected system deviation Δ∥ takes on the valueof zero. In response to changing the power flow direction, i.e., ifbraking power is fed back to current converter SR, then the controldirection of further equalizing controller 32 is reversed. For thispurpose, system deviation Δ∥ for example, of torques-dummy variables{circumflex over (B)}G and BG can be multiplied by sign {circumflex over(P)}_(S) of stator power {circumflex over (P)}_(S), for which purposeprovision is made for a multiplying element 34.

In response to medium and large stator frequencies, it can be expedientto reduce the amplifying effect of controller 32.

This formed system deviation Δ∥ is supplied via a changeover switch 36(position I) to further equalizing controller 32, the control input ofthis changeover switch 36 being connected to an output of a controllingelement 38. On the input side, this controlling element 38 is combinedwith an output of an absolute value generator 40, to whose input isapplied a model stator frequency {circumflex over (ω)}_(S) that isdetected by signal processing 2. The second output (position III) ofchangeover switch 36 is connected via a multiplier 42 in a second inputof an adder 44. The first input of this adder 44 is combined with anoutput of a multiplier 46, which on the input side is connected to theoutput of comparator 24, at whose output system deviation Δ⊥ of torquesactive variables ŴG and WG is applied. On the output side, adder 44 isconnected via a further adder 48 to equalizing controller 22. In eachcase, at the second input of multipliers 42 in 46, a weighting factor K2and K1 is applied. Also sign {circumflex over (ω)}_(r) of estimatedrotor angular frequency {circumflex over (ω)}_(r) is supplied tomultiplier 42 on the output side. For changeover switch 36, a changeoverswitch having a neutral setting is provided.

In addition, further comparator 30 is, on the one hand, combined with adevice 50 for recording peak values and, on the other hand, with adevice 50 for recording the sign of a phase shift angle ξ. A secondinput of this device 50 for detecting the sign is connected to theoutput of equalizing controller 22. On the output side, these devices 50and 50 are, in each case, connected to an input of a multiplier 54,which, on the output side, is combined, using a further changeoverswitch 56 (position I), with an input of adder 48. The control input ofthis further changeover switch 56 is connected to an output of a furthercontrolling element 58, which is joined on the input side to the outputof absolute-value generator 40. The second output of further changeoverswitch 56 (position II) is combined with a further equalizing controller60, which on the output side is connected via an adder 62 to aparameterizing input of signal processing 2. At the second input of thisadder 62, a precontrol value R,0 is applied for the rotor resistanceadaptation. As device 50 for recording peak values, a sample-and-holdelement is provided, device 50 for recording the sign having a phasedetector, a comparator, and an inverter. A block diagram of a device 52of this type is explicitly not provided, since a circuit of this type isknown to the worker skilled in the art. Similarly, further specificembodiments for this device 52 are possible, in accordance with thepresent invention.

As can be seen from this representation, a transducer 64 is connectedupstream of signal processing 2, a value of a nominal stator flux ŜF₀ ofpolyphase machine DM being applied at the input of transducer 64. At theoutput side, a low-frequency modulated flux magnitude signal ŜFM isapplied, which is calculated in accordance with the equation

ŜFM²=[0,72+0,28·sin(ω_(m)·t)]·ŜF₀ ²

or according to the equation

ŜFM=[87,5%+12,5%·sin(ω_(m)·t)]·ŜF₀.

Which equation is used in transducer 64 depends on whether signalprocessing 2 can have the square root extracted or not.

As can also be seen from this representation, changeover switches 36 and56 are actuated as a function of the value of model stator frequency{circumflex over (ω)}_(s). In this context, it is not only the absolutevalue of model stator frequency {circumflex over (ω)}_(s) that is ofinterest. Therefore, from the detected value of model stator frequency{circumflex over (ω)}_(s), a quantity is formed before this value isused for controlling elements 38 and 58. Changeover switches 36 and 56can remain in three positions I, II, and III.

In a model stator frequency {circumflex over (ω)}_(s)=0, changeoverswitch 36 is in position II (neutral setting) and changeover switch 56in position I. In a model stator frequency equivalent to |{circumflexover (ω)}_(s)|>2%, ω₀ giving the type point frequency of polyphasemachine DM, changeover switch 36 is in position I and changeover switch56 is in position II. If the magnitude of model stator frequency{circumflex over (ω)}_(s)>10% ω₀, changeover switch 36 changes toposition II, whereas this changeover switch 36 changes to position III,if for model stator frequency |{circumflex over (ω)}_(s)|>100% ω₀applies. In this value of model stator frequency {circumflex over(ω)}_(s), changeover switch 56 changes to position III (neutralsetting). If changeover switch 36 is in position II or changeover switch56 is in position I or II, then signal processing 2 receives alow-frequency modulated flux magnitude signal ŜFM in place of fluxsetpoint value ŜF. However, the possibility remains of low-frequencymodulated flux magnitude signal, ŜFM for example, being supplied for apredetermined period of time to signal processing 2.

Weighting factors K1 and K2, whose values can be varied between zero andone, change their values as a function of the torque. The value ofweighting factor K1 in response to low torques (<50% M_(breakdown)) isequal to one, and the value of weighting factor K2 is equal to zero. Inresponse to high torques (80% to 100% M_(breakdown)), the value ofweighting factor K1=0 and of weighting factor K2=1. The transitionbetween these two limit cases can, in this context, occur eithercontinuously, with rising loads, or suddenly, after a certain load. Inresponse to a sudden change of weighting factors K1 and K2, it issensible to provide the transition with a hysteresis and thus to avoid aconstant changing between the different controller input variables. Thesign required of rotor angular frequency ω_(r), which also agrees withthe sign of the torque, is detected, in this context, by the completemachine model of signal processing 2.

0≦|{circumflex over (ω)}_(S)|≦2% ω₀:

In this frequency range of model stator frequency {circumflex over(ω)}_(s) changeover switch 36 is in position II and changeover switch 56in position I and low-frequency modulated flux magnitude signal ŜFM issupplied to signal processing 2. The amplitude of this stator fluxmagnitude modulation is, in this context, set so that the torque that isattainable using the lowest value of the modulated stator flux is alwayshigher than the, for example, maximum torque in response to nominalstator flux ŜF₀, which is usually demanded in the traction in responseto these stator frequencies.

Frequency ω_(m) of the flux magnitude modulation must, in this context,be selected to avoid torsion oscillations such that the frequency issignificantly smaller than the lowest mechanical resonance frequency ofthe drive train. The latter is, for example, in the traction forhigh-performance locomotives in the area of 20 Hz. In addition, skineffects are to be avoided in the rotor. For these reasons, isbeneficial, for example, for frequency ω_(m) of the flux magnitudemodulation, to select 5 Hz. Because the cut-off frequency of the torquecontroller of a field-oriented controlling method usually is more thanone order of magnitude higher, the torque nevertheless can be furthercontrolled at its desired value without any problem.

In the case of a difference between the identified rotational velocity{circumflex over (ω)} and rotating velocity ω of polyphase machine DM,on the basis of the stator flux magnitude modulation, a system deviationΔ∥ different from zero of torques-dummy variables {circumflex over (B)}Gand BG arises in the stator currents between model and machine that aredirected parallel to the transformation variable, the stator currents,in this context, corresponding to a periodic quantity that oscillateswith flux magnitude modulation frequency ω_(m). Peak value Δ∥_(max), inthis context, is a function, on the one hand, of the selected amplitudeand the frequency of the stator flux magnitude modulation, but, on theother hand, also significantly of the rotational velocity deviationbetween the machine and the model. The greater the rotational velocityerror, the greater the peak value of system deviation Δ∥ oftorques-dummy variables BG and {circumflex over (B)}G, so that peakvalue Δ∥_(max) delivers supplementary information for the rotationalvelocity measurement. Information concerning the sign of the rotationalvelocity deviation is however not contained here.

A system deviation Δ⊥ of torques-actual variables WG and ŴG, which, inthe event of a rotational velocity difference {circumflex over (ω)}−ωbetween model and machine in flux magnitude modulation, also correspondsto the periodic quantity pulsating at modulation frequency ω_(m), iscompensated for using equalizing controller 22 and is interpreted aspulsating rotating velocity {circumflex over (ω)}. This observed turningcircle frequency {circumflex over (ω)} is then composed generally of asteady component {circumflex over (ω)}_ (median value) and asuperimposed alternating component {circumflex over (ω)}˜.

In FIGS. 3 and 4, the connections in the modulation of the stator fluxmagnitude are depicted. In FIG. 3 machine turning circle frequency ω isgreater than identified median turning circle frequency {circumflex over(ω)}_, whereas in FIG. 4, machine turning circle frequency ω is smallerthan the identified medium turning circle frequency {circumflex over(ω)}_. Median model rotational velocity {circumflex over (ω)}_ is, inthe example, in every case zero. By comparing the representations ofFIG. 3 with FIG. 4, it becomes clear that alternating component{circumflex over (ω)} which is superimposed on identified medianrotational velocity {circumflex over (ω)}_, now lags behind(ω>{circumflex over (ω)}_) and now runs at higher velocity(ω<{circumflex over (ω)}_) than system deviation Δ∥ of torques-dummyvariables BG and {circumflex over (B)}G as a function of presentrotational velocity difference {circumflex over (ω)}−ω between model andmachine. The sign of phase angle ξ between this system deviation Δ∥ andsuperimposed alternating component {circumflex over (ω)}_(˜) of observedturning circle frequency {circumflex over (ω)}, which is measured over aperiod of the stator flux magnitude modulation using known methods, thusdelivers in an unambiguous way the information as to whether theidentified rotational velocity is too large or too small. The followingapplies:

ξ<0,

i.e., alternating component {circumflex over (ω)}_(˜) of observedturning circle frequency {circumflex over (ω)} lags behind systemdeviation Δ∥. From this, it follows that this observed turning circlefrequency {circumflex over (ω)} must be increased.

ξ>0,

i.e., superimposed alternating component {circumflex over (ω)}_(˜) ofobserved turning circle frequency {circumflex over (ω)} runs ahead ofsystem deviation Δ∥. From this, it follows that this observed turningcircle frequency {circumflex over (ω)} must be lowered.

Peak value Δξ_(max) of system deviation Δ∥ of torques-dummy variables{circumflex over (B)}G and BG in is a measure for the existing absolutedeviation between identified rotational velocity {circumflex over (ω)}and machine rotational velocity ω so that with the assistance ofequalizing controller 22 the periodic quantity of this system deviationΔ∥ can be adjusted, the sign of phase angle ξ determining the controldirection.

Through the use of low-frequency modulated flux magnitude signal ŜFM, inthe event of a difference between model turning circle frequency{circumflex over (ω)} and machine turning circle frequency ω, therearises a system deviation Δ∥ of torques-dummy variables {circumflex over(B)}G and BG, which, in this context, basically corresponds to aperiodic quantity oscillating with modulation frequency ω_(m). Themeasurement of peak value Δ∥_(max) which takes place using knownmethods, is described in the depicted example through a sample-and-holdelement. By multiplying by the negated sign of phase shift angle ξbetween system deviation Δ∥ and alternating component {circumflex over(ω)} of identified rotational velocity {circumflex over (ω)},alternating component {circumflex over (ω)}_(˜) being measured usingknown methods over a period of the stator flux magnitude modulation, thecontrol direction is established in an unambiguous manner. Usingchangeover switch 56, this product Δ∥ξ_(m) is additionally used foridentifying the rotational velocity. In this context, the switch overtakes place as a function of the magnitude of model stator frequency{circumflex over (ω)}_(S) which is measured by a complete machine modelof signal processing 2. This variable Δ∥ξ_(m) is additionally used foridentifying the rotational velocity, in that it is added to total systemdeviation ΔΣ with the assistance of adder 48.

2% ω₀<|{circumflex over (ω)}_(S)|<100% ω₀:

In this frequency range of model stator frequency {circumflex over(ω)}_(S), changeover switch 56 is in position II and changeover switch36 is initially in position I and then in position II, and calculatedproduct Δ∥ξ_(m), from peak value Δ∥_(max) of system deviation Δ∥ and thenegated sign, is not additionally used for identifying the rotationalvelocity, but rather for the rotor resistance adaptation. In thiscontext, this value of product Δ∥ξ_(m) using further equalizingcontroller 60 is adjusted to zero in that, in the machine model, acomponent of signal processing 2, the parameter of rotor resistance{circumflex over (R)}_(r) is adjusted. In order that equalizingcontroller 60 should only compensate for its deviations, a precontrol isprovided for. As the precontrol variable, rotor resistance R_(r0) ofpolyphase machine DM is provided for. As a result of this rotorresistance adaptation, rotational velocity ω is no longer identifiederroneously.

|{circumflex over (ω)}_(S)|<100% ω₀

In this frequency range of model stator frequency {circumflex over(ω)}_(S), also known as the field weakening range, changeover switchesis 36 and 56 are each in position III and equalizing controller 22 issupplied with detected system deviation Δ∥ in addition to systemdeviation Δ⊥. In the case of loading, in response to high torques, theinput signal of equalizing controller 22 is ended, which prevents thecontrolled system gain from becoming zero, and no positive feedback canarise from the negative feedback. As a result, it is assured that in thevicinity of the breakdown torque, rotational velocity ω of polyphasemachine DM can clearly be identified. The suppression of individualsystem deviations Δ⊥ and Δ∥ occurs with the assistance of weightingfactors K1 and K2 which can take on the values of zero and one. For acontinuous (stepless) transition from one system deviation Δ⊥ to anothersystem deviation Δ∥, as input signal of equalizing controller 22,weighting factors K1 and K2 can change in a contrary sense using aconstant function, in particular a linear function, between the values 1and 0, as well as 0 and 1. In this overlapping range of weightingfactors K1 and K2, total system deviation ΔΣ is used as input signal ofequalizing controller 22.

As a result of this method according to the present invention, there isnow a method with which rotational velocity ω of a transducerlesspolyphase machine DM, that is operated in a field-oriented manner, canbe identified in the overall rotational velocity range, so that there isno need for a rotational velocity measurement even in high-value drives.In addition, breakdown protection is assured. Additionally, theparameters (stator and rotor resistance) of polyphase machine DM, whichare a function of the operating and working points, are reproducedon-line in the machine model, so that a correct identification of therotational velocity is made possible.

What is claimed is:
 1. A method for determining a rotation velocity of a transducerless polyphase machine that in operating in a field-oriented manner, comprising the steps of: determining a stator-current model space vector and a conjugate complex reference space vector using a complete machine model as a function of a flux setpoint value, a torque setpoint value, an intermediate circuit DC voltage value, measured power converter output voltage values, and system parameters; multiplying each of a measured stator-current model space vector and the determined stator-current model space vector by the conjugate complex reference space vector to form a first product and a second product, respectively; forming a first system deviation of imaginary components by comparing an imaginary component of the first product to an imaginary component of the second product; forming a second system deviation of real components by comparing a real component of the first product to a real component of the second product; weighting the first system deviation of the real components using a sign of a model rotor angular frequency of the polyphase machine and a first factor; weighting the second system deviation of the imaginary components using a second factor; adding the weighted first system deviation of the real components to the weighted second system deviation of the imaginary components to form a weighted total system deviation; adjusting the model rotor angular velocity as a function of the weighted total system deviation so that the weighted total system deviation becomes zero; and determining the rotational velocity of the transducerless polyphase machine as a function of the model rotor angular velocity.
 2. The method according to claim 1, further comprising the steps of: providing a low-frequency modulated flux magnitude signal as the flux magnitude setpoint value; measuring a sign of a phase shift between the model rotor angular velocity and the second system deviation of the real components; measuring a peak value of the second system deviation of real components; negating the sign of the phase shift between the model rotor angular velocity and the second system deviation of the real components; multiplying the peak value with the negated sign to form a third product; and superimposing the third product on the measured total system deviation.
 3. The method according to claim 1, further comprising the steps of: providing a low-frequency modulated flux magnitude signal as the flux magnitude setpoint value; measuring a sign of a phase shift angle between the model rotor angular velocity and the second system deviation of the real components; measuring a peak value of the second system deviation of the real components; negating the sign of the phase shift between the model rotor angular velocity and the second system deviation of the real components; multiplying the peak value with the negated sign to form a third product; and adjusting a rotor resistance parameter of the system parameters using the third product so that the third product becomes zero.
 4. The method according to claim 1, further comprising the step of: varying the first factor and the second factor between zero and one and contrary to each other.
 5. The method according to claim 1, wherein, in a field weakening range of the polyphase machine, the first factor is set to zero for high torques, and the second factor is set to one.
 6. The method according to claim 4, wherein the varying step includes varying the first factor and the second factor as a function of magnitudes of a model stator frequency of the polyphase machine and the estimated value of the rotor angular frequency.
 7. The method according to claim 2, wherein the adding step is performed in response to a magnitude of a model stator frequency of the polyphase machine that is one of greater than and equal to two percent of a type point frequency of the polyphase machine.
 8. The method according to claim 3, wherein the adjusting the rotor resistance parameter step is performed in response to a magnitude of a model stator frequency of the polyphase machine that is one or greater than and equal to two percent of a type point frequency of the polyphase machine.
 9. The method according to claim 7, wherein the low-frequency modulated flux magnitude signal is turned on only for a selected time.
 10. A device for determining a rotation velocity of a transducerless polyphase machine that in operating in a field-oriented manner, comprising: a signal processor including a complete machine model of the polyphase machine, a closed-loop controller, and a modulator; a first multiplier and a second multiplier, the signal processor applying a conjugate complex reference space vector to a first input of the first multiplier and a first input of the second multiplier, the signal processor applying a stator-current module space vector to a second input of the first multiplier, a first output of the first multiplier providing an imaginary component of a product formed by the first multiplier and a second output of the first multiplier providing a real component of the product formed by the first multiplier, a first output of the second multiplier providing an imaginary component of a product formed by the second multiplier and a second output of the second multiplier providing a real component of the product formed by the second multiplier; a current measuring element, a second input of the second multiplier being coupled to an output of the current measuring element using a resolver; a first comparator, the first output of the first multiplier and the first output of the second multiplier being coupled by the first comparator; a second comparator, the second output of the first multiplier and the first output of the second multiplier being coupled by the second comparator; a first adder, an output of the first comparator being coupled to a first input of the first adder via a third multiplier, an output of the second comparator being coupled to a second input of the first adder via a fourth multiplier; a first equalizing controller, an output of the first adder providing, via the first equalizing controller, a model rotor angular velocity to a first parameter input of the signal processor; a first changeover switch coupled upstream of the fourth multiplier; and an absolute-value generator having a downstream controlling element for the first changeover switch, the signal processor applying a model stator frequency to the absolute value generator, the absolute value generator coupling the model stator frequency to a controller input of the first changeover switch.
 11. The device according to claim 10, further comprising: a transducer coupled upstream of the signal processor; a peak value detector, the output of the second comparator being coupled to the peak value detector; a sign of phase shift angle detector, the output of the second comparator being coupled to the sign of phase shift angle detector; a fifth multiplier, an output of the peak value detector and an output of the sign of phase shift angle detector coupled to inputs of the fifth multiplier; a second adder, a first input of the second adder coupled to the output of the first adder; a second changeover switch, an output of the fifth multiplier coupled to a second input of the second adder via the second changeover switch, an output of the second adder coupled to the first equalizing controller, a control input of the second changeover switch being coupled via a further controlling element with the output of the absolute-value generator.
 12. The device according to claim 10, further comprising: a transducer coupled upstream of the signal processor; a peak value detector, the output of the second comparator coupled to the peak value detector; a sign of phase shift angle detector, the output of the second comparator coupled to the sign of phase shift angle detector; a third multiplier, an output of the peak value detector and an output of the sign of phase shift detector coupled to inputs of the third multiplier; a second changeover switch; a second equalizing controller, an output of the third multiplier being coupled to the second equalizing controller via the second changeover switch; and a further controlling element, a control input of the second changeover switch coupled via the further controlling input to the output of the absolute-value generator.
 13. The device according to claim 12, further comprising: a second adder, an output of the second equalizing controller being coupled to a first input of the second adder, a second input of the second adder receiving a precontrol value for rotor resistance adaptation, an output of the second adder providing a model rotor resistance to a second parameter input of the signal processor.
 14. The device according to claim 10, wherein the peak value detector includes a sample-and-hold element.
 15. The device according to claim 10, wherein the signal of the phase shift angle detector includes a phase detector, a third comparator, and an inverter.
 16. The device according to claim 10, wherein the first changeover switch has a neutral setting.
 17. The device according to claim 10, wherein the downstream controlling element includes a third comparator. 